IXMS150PSI datasheet(6/10 Pages) IXYS

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IXMS 150

the H-bridge that must be properly

filtered if the system is to operate with

the desired degree of precision.

This presents a significant engineering

challenge that has been solved by

IXYSs design team. Using proprietary

analog and digital signal processing

techniques, IXYS has developed a

control system that measures the true

average phase currents. Requiring only

one sense resistor per H-bridge, this

technique avoids errors due to mis-

matches in charge/discharge currents

associated with using one sense resi-

stor on each leg of the H-bridge. This

improves system performance as well

as minimizing component count. The

sense resistor for each H-bridge should

be selected based on the required peak

motor current:

S = 0.625 V/Impk

(9)

The voltage developed across this

resistor is then applied to the corres-

ponding sense input for each H-bridge.

Negative bias Generator

One of todays cost cutting trends is to

minimize the number of power supplies,

implying single supply operation for the

control section. Yet the current feed-

back and reference inputs are bipolar

signals. Level shifting has been used

for the reference input in the past, but

that can not be easily done for the

feedback signal without impacting

accuracy or efficiency. In practice one

finds that in order to generate true zero

voltage having low impedance drive

there must be a negative power supply.

Otherwise there will be a tradeoff

sacrificing accuracy for simpler system

design.

For these reasons the approach selec-

ted by IXYS was different. Taking

advantage of our CMOS design, we

opted to build into the chip a negative

bias generator. This does put stringent

demands on noise coupling but results

in the most flexible system having the

highest possible accuracy. The built in

charge pump circuit requires two capa-

citors and two diodes to be added

externally. The recommended compo-

nent values for an oscillator frequency

of 100 kHz are given below.

C1 = 0.047 µF

C2 = 100 µF

D1 = D2 = 1N4148

Note: V

BB = -(VDD/5)

in torque value errors but no positioning

errors. The question is, what is the

upper bound on the current errors in

order to keep the position error within

some given angle ∆θ.

Referring to Fig. 3, assume the required

currents i

A, iB are given by Equations

(1), (2) respectively such that their

vector sum points to position P. Let the

phase currents vary by a small amount

such that their vector sum lies within a

circle centered at point P and having

the radius i, as indicated in Fig. 3.

Fig. 3 Effect of Current Errors on

Position

If follows that the worst case position

error occurs for the cases where the

vector sum is tangent to the circle such

as point P

1, at which:

tan (∆ θ) = ∆ i/l

(6)

For instance, to keep position error to

less than 1% of a full step, the electrical

angular error would be:

∆ θ = 0.01 90° = 0.9°

(7)

This is assuming there are 90 electrical

degrees for a full step. Therefore total

current error must be:

i/I

0 = tan (∆θ) = 0.016 or 1.6 %)

(8)

Thus the current error must be kept to

less than 1.6 % of full scale or peak

current at each phase for 1 % maximum

position error. This upper bound on

error includes all error sources such as

zero offset errors and full scale matching

errors. Another interesting observation

is that in the vicinity of a full step (i.e.,

θe = 0), the phase having the bigger

impact on position error is the one

carrying the smaller current through it.

This has a strong impact on input

waveform generation.

Input waveform generation

It has been shown that the two input

signals, VIN

A and VINB, are sinusoidal

and 90° out of phase. This may be

accomplished by using two look-up

Fig. 4 Simple Reference Waveform

Generator

tables stored in ROM and two DACs

per Fig. 4. An up/down counter may be

used to generate the appropriate

address locations for the ROMs and the

data outputs used to control the DACs.

The user then need only supply up or

down pulses to the counter to control

the IXMS150 and hence the motor.

In higher performance systems a

microprocessor may be used in place

of the counter and the ROMs. The

micro can perform the look-up function

and calculate the appropriate system

responses, velocity profiles, etc.

necessary for total system operation.

An example of this configuration is

shown in Fig. 5.

Current Sensing Considerations

Most commercially available monolithic

PWM controllers monitor and control

the peak of the phase current by com-

paring the voltage across the sense

resistor with a ramp voltage. This

approach assumes that the ripple

current is fixed in amplitude. Results

shown later clearly indicate the varia-

tion of the ripple current with frequency.

But even in fixed frequency systems

the ripple current is directly proportional

to the motor supply voltage and to the

back EMF voltage of the motor. Ripple

current is not insignificant compared to

the full scale current and therefore

cannot be neglected in a precision

system. In addition, there are transients

associated with the turn on and turn off

characteristics of the power devices in

Fig. 5 Microprocessor Based

Referenced Waveform Generator

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